Power

RCD Clamp Snubber Calculator — Flyback Leakage-Inductance Clamp

Design the RCD clamp that limits a flyback converter's drain voltage spike: clamp power including the reflected-voltage factor, the clamp resistor and capacitor, the peak drain voltage and the diode's reverse-voltage rating, from the leakage inductance, peak current and clamp voltage.

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Formula Reference

P_{sn} = \tfrac{1}{2} L_{lk} I_{pk}^2 f_s \frac{V_{sn}}{V_{sn} - nV_o},\quad R_{sn} = \frac{V_{sn}^2}{P_{sn}},\quad C_{sn} = \frac{V_{sn}}{\Delta V_{sn} R_{sn} f_s},\quad V_{ds,pk} = V_{in,max} + V_{sn}

Ref: Koo, Design Guidelines for RCD Snubber of Flyback Converters, Fairchild AN-4147, Rev. 1.1.0 (2006), Eqs. (2)–(5)

L_{lk}— Primary leakage inductance (µH)

I_{pk}— Peak primary current at turn-off (A)

f_s— Switching frequency (kHz)

nV_o— Reflected output voltage (V)

V_{sn}, \Delta V_{sn}— Clamp voltage and its ripple (V)

P_{sn}, R_{sn}, C_{sn}— Clamp power, resistor and capacitor (W, kΩ, nF)

Formula

P_{sn} = \tfrac{1}{2} L_{lk} I_{pk}^2 f_s \frac{V_{sn}}{V_{sn} - nV_o},\quad R_{sn} = \frac{V_{sn}^2}{P_{sn}},\quad C_{sn} = \frac{V_{sn}}{\Delta V_{sn} R_{sn} f_s},\quad V_{ds,pk} = V_{in,max} + V_{sn}

Reference: Koo, Design Guidelines for RCD Snubber of Flyback Converters, Fairchild AN-4147, Rev. 1.1.0 (2006), Eqs. (2)–(5)

L_{lk}— Primary leakage inductance (µH)

I_{pk}— Peak primary current at turn-off (A)

f_s— Switching frequency (kHz)

nV_o— Reflected output voltage (V)

V_{sn}, \Delta V_{sn}— Clamp voltage and its ripple (V)

P_{sn}, R_{sn}, C_{sn}— Clamp power, resistor and capacitor (W, kΩ, nF)

V_{ds,pk}— Peak drain voltage, which the diode must also block (V)

How It Works

In a flyback converter not all of the primary's flux links the secondary. The part that does not is the leakage inductance $L_{lk}$ , and when the MOSFET turns off, the current in it has nowhere to go: it charges the MOSFET's output capacitance and the drain voltage spikes far above the input plus the reflected output voltage $V_{in} + nV_o$ . An RCD clamp catches that spike. A diode from the drain charges a capacitor that sits at a clamp voltage $V_{sn}$ above the input rail, and a resistor across the capacitor bleeds off the energy each cycle.

Clamp power

While the clamp conducts, the secondary holds the magnetising inductance at $nV_o$ , so $V_{sn} - nV_o$ falls across the leakage inductance and its current falls from the peak $I_{pk}$ to zero in

t_s = \frac{L_{lk} I_{pk}}{V_{sn} - nV_o}

The clamp absorbs $V_{sn}I_{pk}t_s/2$ per cycle, which gives the clamp power of Fairchild's application note AN-4147 (Koo, Rev. 1.1.0, 2006):

P_{sn} = \tfrac{1}{2} L_{lk} I_{pk}^2 f_s \frac{V_{sn}}{V_{sn} - nV_o}

The first part is the energy stored in the leakage inductance. The factor $V_{sn}/(V_{sn} - nV_o)$ is the extra energy the transformer pushes into the clamp while the leakage current falls: it is 2 when the clamp sits at twice the reflected voltage, and it grows without limit as $V_{sn}$ approaches $nV_o$ . At or below $nV_o$ the clamp would conduct continuously and carry the output power, so the calculator refuses it. AN-4147 recommends $V_{sn}$ of 2 to 2.5 times $nV_o$ .

Resistor, capacitor and voltage stress

The resistor dissipates $V_{sn}^2/R_{sn}$ , so $R_{sn} = V_{sn}^2/P_{sn}$ . The capacitor is sized for the allowed ripple $\Delta V_{sn}$ , usually 5 to 10% of $V_{sn}$ :

C_{sn} = \frac{V_{sn}}{\Delta V_{sn} R_{sn} f_s}

With the clamp returned to the input rail, the drain peaks at $V_{in,max} + V_{sn}$ , and the diode blocks the same voltage while the MOSFET is on. AN-4147 asks for the steady-state peak to stay below 80% of the MOSFET's rated voltage, and for an ultrafast diode rated above it.

Validity

The equations assume the clamp capacitor's voltage stays constant over a switching period, which AN-4147 takes as holding for ripple up to 10%; above that the calculator warns. They also assume the leakage current falls to zero within the switching period, which it cannot do when $t_s$ is longer than the period; that is flagged too. The values are still shown in both cases. Leakage inductance and peak current should be taken at the design point AN-4147 uses, minimum input voltage and full load, where the clamp power is highest.

Worked Example

Problem: An off-line flyback converter for 85 to 265 V AC switches at 65 kHz. Its maximum DC input is 375 V, the reflected output voltage is 100 V, the measured primary leakage inductance is 8 µH and the peak primary current at minimum input and full load is 1.2 A. Clamp at 220 V (2.2 times the reflected voltage) with 8% ripple.

Step 1 - Leakage energy alone: ½ × 8 µH × (1.2 A)² × 65 kHz = 0.374 W

Step 2 - Reflected-voltage factor and clamp power: 220 / (220 − 100) = 1.833 P_sn = 0.374 W × 1.833 = 0.686 W

Step 3 - Clamp resistor: R_sn = (220 V)² / 0.686 W = 70.5 kΩ

Step 4 - Clamp capacitor: ΔV_sn = 8% × 220 V = 17.6 V C_sn = 220 / (17.6 × 70.5 kΩ × 65 kHz) = 2.73 nF

Step 5 - Voltage stress: V_ds,pk = 375 V + 220 V = 595 V, so the diode must block 595 V as well.

Step 6 - Diode conduction time: t_s = 8 µH × 1.2 A / (220 − 100) V = 80 ns, far inside the 15.4 µs period.

An 800 V MOSFET keeps 595 V within AN-4147's 80% rule, and the clamp diode should be an ultrafast part rated 800 V or more. A clamp at 150 V would cost 1.12 W (factor 3); one at 250 V costs 0.624 W but lifts the drain to 625 V.

Practical Tips

✓Start with V_sn at 2 to 2.5 times the reflected voltage, as AN-4147 recommends, then trade clamp loss against drain voltage.
✓Rate the clamp resistor at least twice its dissipation, and the capacitor for V_sn with margin; a film or high-voltage C0G capacitor holds its value at that voltage.
✓Reduce the leakage inductance first: interleaved windings and good coupling lower the clamp power more than any choice of R and C.
✓Verify on the bench: the steady-state drain peak should stay below 80% of the MOSFET's rating, and the start-up peak below 90% (AN-4147).
✓Export the BOM or KiCad schematic to get the diode, resistor and capacitor sized for current, power and voltage, with the diode from the drain and R and C from the clamp node to the input rail.

Common Mistakes

✗Sizing the clamp resistor from the leakage energy alone, ½·L·I²·f. The clamp also absorbs the energy the transformer drives into it while the leakage current falls; at twice the reflected voltage that doubles the power.
✗Setting the clamp voltage barely above the reflected voltage. The loss factor V_sn/(V_sn − nV_o) then becomes very large, and the clamp starts to carry output power.
✗Using a standard-recovery rectifier such as a 1N4007. Its slow reverse recovery lets the clamp capacitor discharge back into the drain each cycle; use an ultrafast diode rated above the MOSFET's breakdown voltage.
✗Measuring leakage inductance with the secondary open. It must be measured at the primary with every secondary winding shorted.
✗Checking the drain voltage only at minimum input. The peak drain voltage is V_in,max + V_sn and must be checked at the highest input voltage.

Frequently Asked Questions

Why is the clamp power more than the leakage energy?

While the clamp conducts, the transformer's reflected voltage keeps pushing current into it until the leakage current has fallen to zero. That adds energy in proportion to nV_o, which the factor V_sn/(V_sn − nV_o) accounts for. At a clamp voltage of twice the reflected voltage the clamp dissipates twice the leakage energy.

What clamp voltage should I choose?

AN-4147 suggests 2 to 2.5 times the reflected output voltage. Lower clamps dissipate more, because the loss factor rises steeply towards V_sn = nV_o; higher clamps dissipate less but raise the peak drain voltage, V_in,max + V_sn.

What reverse voltage must the clamp diode withstand?

While the MOSFET is on, its drain is near ground and the clamp node sits at V_in + V_sn, so the diode blocks V_in,max + V_sn, the same as the peak drain voltage. AN-4147 recommends rating it above the MOSFET's breakdown voltage, and an ultrafast type.

How large should the clamp capacitor be?

Large enough that its voltage changes little over one cycle: C_sn = V_sn / (ΔV_sn·R_sn·f_s), with ΔV_sn usually 5 to 10% of V_sn. A larger capacitor gives less ripple but takes longer to settle at start-up.

Can I use an RC snubber instead?

An RC snubber damps ringing but dissipates C·V²·f whatever happens; across a flyback's drain, with a capacitor big enough to absorb the leakage energy, that loss is far higher than a clamp's. The two are often combined: an RCD clamp for the spike and a small RC snubber for the residual ringing.

Methodology & References

References

Design Guidelines for RCD Snubber of Flyback Converters — Gwan-Bon Koo, Fairchild Semiconductor Application Note AN-4147, Rev. 1.1.0 (2006) — Eqs. (1)–(5), pp. 2–3: conduction time, clamp power, resistor and capacitor; pp. 3–4: diode choice, the 80% drain-voltage rule and the FSDM311 design example, Eqs. (9)–(11)
Snubber Circuits: Theory, Design and Application — Philip C. Todd, Unitrode Corporation (May 1993), TI literature SLUP100 — p. 2-7: the RCD voltage snubber in clamp mode, its resistor returned to the input bus, dissipating the leakage energy ½·L·I²·f plus the energy from the reflected voltage

Reproduces AN-4147's design example (14 kΩ, 1.6 W and, by its Eq. (11), 10.66 nF; 525 V predicted against the 524 V measured in its Figure 7); a clamp at twice the reflected voltage doubles the leakage energy, and a large clamp voltage tends to the leakage energy alone.

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